Digitally controlled integrated dc-dc converter with transient suppression

ABSTRACT

A fully integrated DC-DC converter utilizes digitally controlled dual output stages to achieve fast load transient recovery is presented. The DC-DC converter includes a main converter output stage connected in parallel with an auxiliary output stage. The main output stage is responsible for steady-state operation and is designed to achieve high conversion efficiency using large inductor and power transistors with low on-resistance. The auxiliary stage is responsible for transient suppression and is only active when a load transient occurs. The auxiliary output stage performs well with inductor and power transistors much smaller than those of the main switching stage and thus achieves well balanced power conversion efficiency and dynamic performance with a much smaller area penalty than previously described dual-output-stage converters.

FIELD OF THE INVENTION

The present invention relates in general to digitally controlledintegrated DC-DC converters. More specifically, the present inventionrelates to a digitally controlled integrated DC-DC converter thatincorporates transient suppression.

BACKGROUND

Digitally controlled DC-DC converters have shown more flexibility overtheir analog counterparts with the introduction of intelligent controltechniques. The intelligent control techniques include, for example, theuse of an auto-tuning system that can tolerate passive componentvariations as described in “Limit-cycle oscillations based auto-tuningsystem for digitally controlled DC-DC power supplies”, by Z. Zhao, A.Prodić, IEEE Trans. Power Electronics, vol. 24, no. 6, pp. 2211-2222,November 2007, the use of a segmented output stage that can dynamicallyadjust the size of the output transistors according to load conditionsto maintain high power conversion efficiency as described in “Adigitally controlled DC-DC converter module with a segmented outputstage for optimized efficiency”, by O. Trescases, W. T. Ng, H. Nishio,M. Edo and T. Kawashima, Proc. Int. Symp. Power Semiconductor Devicesand ICs, June 2006, pp. 373-376, and the one-step dead-time correctionthat can optimize the turn-on and turn-off dead-time for powertransistors on-the-fly as described in “One-step digital dead-timecorrection for DC-DC converters”, by A. Zhao, A. A. Fomani and W. T. Ng,Proc. Applied Power Electronics Conf., February 2010, pp. 132-137. Inaddition, digitally controlled DC-DC converters have the ability toswitch seamlessly between linear and nonlinear operation modes andachieve near-optimal load transient performance as described in “Minimumdeviation digital controller IC for single and two phase DC-DCswitch-mode power supplies”, by A. Radic, Z. Lukic, A. Prodic and R. deNie, Proc. Applied Power Electronics Conf., February 2010, pp. 1-6.

As power supply requirements for microprocessors become more stringent,however, the design of power converters has become more challenging.Point-of-load (POL) DC-DC converters driving modern microprocessors needto provide low output voltage, high output current and good dynamicperformance during load transients, while at the same time maintaininghigh efficiency. Smaller LC filters and higher switching frequency ormultiphase/interleaved structure have been proposed to improve theconverter's transient performance. See, for example, “CriticalInductance in Voltage Regulator Modules”, P. L. Wong, F. C. Lee, Px Xuand K. Yao, IEEE Transaction on Power Electronics, Vol. 17, No. 4, July2002, pp. 485-492. However, these solutions usually suffer fromefficiency degradation. In order to address this problem, an additionalpower output stage with much smaller filter inductance has been added tothe main converter to reduce the output voltage overshoot withoutdeteriorating the steady-state efficiency. See, for example, “A fasttransient recovery module for DC-DC converters”, by P. J. Liu, H. J.Chiu, Y. K. Lo, and Y.-J. E. Chen, IEEE Trans. Industrial Electronics,vol. 56, no. 7, pp. 2522-2529, July 2009, the content of which isincorporated herein by reference. However, depending on theimplementation, the use of auxiliary stages can take up valuable space.

A digitally controlled transient suppression method involving anauxiliary output stage connected in parallel with the main output stagehas been proposed in “A Digitally Controlled Transient SuppressionMethod for Integrated DC-DC Converters”, K. NG, J. Wang, and W. T. Ng,IEEE International Conference on Electron Devices and Solid-StateCircuits (EDSSC), December 2008, the content of which is incorporatedherein by reference. In the digitally controlled transient suppressionmethod, a capacitor charge balance principle is applied to bring theoutput voltage back to within a tolerable window in a single switchingcycle, which in turn results in a very short recovery time. Theauxiliary stage is used to assist the sinking or sourcing of the loadcurrent, which helps to restore the output voltage quickly when loadtransient occurs. It is desirable that that the auxiliary powertransistors be much smaller than those in the main output stage. As aresult, the total area required for dual output stages does not impose asignificant overhead when compared to converters with a conventionalsingle phase output stage, thereby making the auxiliary stage a methodviable for integration.

In view of the above, it would therefore be desirable to provide a fullyintegrated DC-DC converter that achieves a fast load transient recovery,while at the same time achieving well balanced power conversionefficiency and dynamic performance without a significant area penalty.

SUMMARY OF THE INVENTION

A fully integrated DC-DC converter utilizes digitally controlled dualoutput stages to achieve fast load transient recovery is presented. TheDC-DC converter includes a main converter output stage connected inparallel with an auxiliary output stage. The main output stage isresponsible for steady-state operation and is designed to achieve highconversion efficiency using large inductor and power transistors withlow on-resistance. The auxiliary stage is responsible for transientsuppression and is only active when a load transient occurs. Theauxiliary output stage performs well with inductor and power transistorsmuch smaller than those of the main switching stage and thus achieveswell balanced power conversion efficiency and dynamic performance with amuch smaller area penalty than previously described dual-output-stageconverters.

More specifically, in a preferred embodiment of the invention, the DC-DCconverter includes a main converter output stage, an auxiliary converteroutput stage connected in parallel to the main converter output stage,and a digital controller coupled to the main converter output stage andthe auxiliary converter output stage. The digital controller operates ina linear controller mode when no load transient is present and operatesin a non-linear controller mode when a load transient is detected. Thedigital controller senses an output voltage slew-rate when the loadtransient is detected and determines a duty-ratio prediction that isapplied in the linear controller mode to ensure smooth transition fromthe nonlinear controller mode back to the linear controller mode. Thelinear control mode is preferably a proportional-integral-derivative(PID) control mode.

The main converter output stage and the auxiliary converter output stageeach include a high side switch, a low side switch and an inductor. Theswitches of both stages are coupled to the digital controller to enablethe digital controller to control the operation of both stages. When noload transient is detected and the DC-DC converter is operated in asteady state condition, the digital controller disables the auxiliaryconverter output stage.

An on-resistance of auxiliary converter output stage power transistorsprovided in the auxiliary converter output stage is such that theauxiliary converter output stage active time is less than the mainconverter stage recovery time. This allows the size of the auxiliaryconverter output stage to be minimized.

In operation, the digital controller functions in a linear control modewhen no load transient is detected by the digital controller in order tomaintain the DC-DC converter in a steady state condition. The digitalcontroller is then activated to operate in a non-linear control modewhen a load transient is detected. The digital controller then performsload-step sensing in order to generate the necessary switching commandsfor the main converter output stage and the auxiliary converter outputstage. In addition, the digital controller performs a duty-ratioprediction and applies it to the linear control mode to ensure a smoothtransition from the non-linear control mode to the linear control mode.Further, a blocking state is preferably utilized until the outputvoltage of the converter settles to steady state.

These and other features, advantages, modifications and embodiments willbecome apparent to one skilled in the art after review of the followingdetailed description of the preferred embodiments of the invention andthe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described with reference to certain preferredembodiments thereof and the accompany drawings, wherein:

FIG. 1 is a schematic block diagram of a DC-DC converter incorporatingdigitally controlled dual output stages in accordance with the presentinvention;

FIGS. 2A-2D respectively illustrate theoretical waveforms for loadcurrent (I_(out)), main stage inductor current (I_(LM)), auxiliary stageinductor current (I_(LA)) and output voltage variation (V_(out)-V_(ref))under heavy-to-light load transient for the DC-DC converter illustratedin FIG. 1;

FIG. 3 is a state diagram illustrating the operation of a digitalcontroller incorporated into the DC-DC converter illustrated in FIG. 1;

FIG. 4 is a graph illustrating finding the range of inductanceL_(M)/L_(A);

FIGS. 5A-5C illustrate theoretical waveforms of load current (I_(out)),main stage inductor current (I_(LM)), auxiliary stage inductor current(I_(LA)) under heavy-to-light load transient taking into account theon-resistances of the auxiliary converter output stage switches for theDC-DC converter illustrated in FIG. 1;

FIG. 6 is a graph illustrating dynamic performance of converters usingauxiliary switches with different on resistances;

FIG. 7 is a graph illustrating transient current in the auxiliaryswitches with different on-resistances;

FIGS. 8 and 9 are graphs illustrating current in the main stageinductor;

FIG. 10 is a graph illustrating a comparison of the active time forauxiliary output stages with different on-resistances;

FIG. 11 is a sample layout of a typical output stage showing therelative sizes of the main and auxiliary output stages;

FIG. 12 illustrates the sensing of the output voltage slew-rate of theDC-DC converter illustrated in FIG. 1 in order to determine a duty-cycleprediction;

FIG. 13 illustrates dynamic response under a 2.25 A-to-0.25 A loadtransient using a conventional single-stage device; and

FIG. 14 illustrates dynamic response under a 2.25 A-to-0.25 A loadtransient using a dual output stages in accordance with the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS OF THE INVENTION

A fully integrated DC-DC converter 10 that utilizes digitally controlleddual output stages to achieve fast load transient recovery in accordancewith the present invention is shown in FIG. 1. As shown in FIG. 1, theDC-DC converter 10 includes a main converter output stage 12 connectedin parallel with an auxiliary converter output stage 14. A digitalcontroller 16 is provided to control the overall operation of the DC-DCconverter 10. The main converter output stage 12 is responsible forsteady-state operation and is designed to achieve high conversionefficiency using large inductor and power transistors with lowon-resistance. The auxiliary converter output stage 14 is responsiblefor transient suppression, is only active when a load transient occurs,and is implemented with a smaller inductor so that is can source or sinkcurrent with a high slew-rate to quickly recover the output voltage toits steady-state value. The digital controller 16 generates switchingcommands for the main converter output stage 12 and the auxiliaryconverter output stage 14 using the charge balance principle describedin “An optimal control method for buck converters using a practicalcapacitor charge balance technique”, by E. Meyer, Z. Zhang, and Y. F.Liu, IEEE Trans. Power Electronics, vol. 23, no. 4, pp. 1802-1812, July2008, the content of which is incorporated herein by reference, suchthat the amount of charge released through the auxiliary converteroutput stage 14 equals the excess charge injected by the main converteroutput stage 12.

An example of the theoretical current waveforms for both the mainconverter output stage 12 and the auxiliary output stage 14 during aheavy-to-light load transient are as illustrated in FIGS. 2A-2D. It isassumed that the DC-DC converter 10 reacts immediately to the loadcurrent step. After a load transient with magnitude ΔI_(out) isdetected, the current I_(LM) of the main converter output stage 12inductor decreases with a constant slope of k₁ until it reaches thelevel of the new steady-state load current by the end of T_(R). In themean time, during t_(on) the auxiliary converter output stage 14inductor draws current I_(LA) from the output capacitor at slope k₂,then ramps back to zero with a slope of k₃ through t_(off). The threecontrol parameters: main stage recovery time (T_(R)), auxiliary stageon-time (t_(on)) and auxiliary stage off-time (t_(off)) are determinedby the digital controller 16 such that the capacitor charge releasedthrough the auxiliary converter output stage 14 equals the excess chargeinjected by the main converter output stage 12. By equating the area ofthe two shaded triangles in FIGS. 2B and 2C, these control parametersare calculated as:

$\begin{matrix}{{T_{R} = \frac{\Delta \; I_{out}}{k_{1}}},} & (1) \\{{t_{on} = {\Delta \; I_{out}\sqrt{\frac{\left( {V_{in} - V_{out}} \right)}{V_{in}k_{1}k_{2}}}}},} & (2) \\{{t_{off} = {\frac{k_{2}}{k_{3}}t_{on}}},{where}} & (3) \\{{k_{1} = \frac{V_{out}}{L_{M}}},} & (4) \\{{k_{2} = \frac{V_{out}}{L_{A}}},} & (5) \\{k_{3} = {\frac{V_{in} - V_{out}}{L_{A}}.}} & (6)\end{matrix}$

The parameters V_(in) and V_(out) are the input and output voltages ofthe DC-DC converter 10, while L_(M) and L_(A) are the inductances in themain converter output stage 12 and the auxiliary output stage 14respectively. For any given design, these values as well as k₁, k₂, andk₃ can be seen as known constants. The only unknown parameter needed tocalculate T_(R), t_(on), and t_(off) is the real-time load current stepΔI_(out).

The digital controller 16 is implemented to generate the switchingcommands for the main converter output stage 12 and the auxiliaryconverter output stage 14 during both a steady state condition ofoperation and a transient recovery process. A state diagram illustratinga preferred operation of the digital controller 16 is shown in FIG. 3.After power up, the digital processor 16 soft starts when the outputvoltage gradually rises to its targeted value and settles in steadystate condition under a conventional PID control mode. Only the mainconverter output stage 12 is active during this period. When a loadtransient is detected by the digital controller 16, the PID control modeis suspended and the system enters a non-linear control mode where boththe main converter output stage 12 and auxiliary converter output stage14 switch to perform transient recovery. The control parameters TR, ton,and toff for different load step Δ/out are calculated in advance fromEquations (1)-(3) and programmed into a look-up table provided in thedigital controller 16 (or memory associated therewith). The digitalcontroller 16 simply uses sensed Δ/out as an index to determine TR, ton,toff and to generates the switching commands. By the end of transientrecovery process, a new steady-state duty-ratio after transient ispredicted in accordance with Δ/out. The new duty-ratio is then appliedto the PID control mode when it reactivates to achieve seamlesstransition back to PID linear control. A short period of transientblocking state is introduced afterwards when the system is forced tooperate with the PID control mode to prevent malfunctioning of thenon-linear control until the output voltage fully settles. The blockingstate ends when the sensed error signal of the output voltage staysaround zero for a predetermined number of switching cycles. Due to theloss elements in the power path, the actual steady-state duty-ratio seenat the main converter output stage's switching node changes withdifferent load current even if the ratio of output voltage over inputvoltage remains the same.

The ratio of inductance L_(M)/L_(A) is preferably determined such thatduring transient recovery the induced output voltage undershootΔV_(Undershoot) is always less than or equal to the output voltageΔV_(Overshoot) overshoot for ideal heavy-to-light load transients witharbitrary magnitude, as shown in FIG. 2D. For the converter underdiscussion, ΔV_(Overshoot) and ΔV_(Undershoot) can be derived as:

$\begin{matrix}{{{\Delta \; V_{Overshoot}} = {\frac{\Delta \; I_{out}^{2}}{2C_{out}} \cdot \frac{1}{k_{1} + k_{2}}}},} & (7) \\{{{\Delta \; V_{Undershoot}} = {\frac{\Delta \; I_{out}^{2}}{2C_{out}} \cdot \frac{k_{1}}{1 - \frac{k_{1}}{k_{3}}} \cdot \left\lbrack {\frac{1}{k_{1}} - \sqrt{\frac{1}{\left( {1 - D} \right)k_{1}k_{2}}}} \right\rbrack^{2}}},} & (8) \\{{{{where}\mspace{14mu} D} = \frac{V_{out}}{V_{in}}},} & (9)\end{matrix}$

C_(out) is the converter's output capacitance and k₁˜k₃ are defined asin Equation (4)˜(6). assuming

$\begin{matrix}{{\frac{L_{M}}{L_{A}} = x},} & (10)\end{matrix}$

and letting

ΔV _(Undershoot) ≦ΔV _(Overshoot)

by manipulating Equation (11), the following inequality is obtained:

$\begin{matrix}{{\frac{1 + x}{1 - {\frac{D}{1 - D} \cdot \frac{1}{x}}} \cdot \left\lbrack {1 - \sqrt{\frac{1}{\left( {1 - D} \right)x}}} \right\rbrack^{2}} \leq 1} & (12)\end{matrix}$

The left side of Equation (12) can be plotted in relation to x for anygiven combination of V_(in) and V_(out). The range of x that satisfiesEquation (12) can be easily located graphically as shown in FIG. 4. Alarger x is preferable since it results in a smaller auxiliary converteroutput stage inductor.

The size of the power transistors of the auxiliary converter outputstage 14 is determined by analyzing the influence of theiron-resistances R_(on) _(—) _(A). The theoretical current and outputvoltage waveforms taking into account R_(on) _(—) _(A) during aheavy-to-light transient are as illustrated in FIGS. 5A-5C.On-resistances of the power transistors of the main converter outputstage 12 HS_(main) and LS_(main) are small due to the requirement ofhigh steady-state efficiency. Thus its influence can be ignored andEquations (1) & (4) are viable. The voltage difference between V_(in)and V_(out) is much larger than the voltage drop across the high-sideauxiliary transistor HS_(Aux). Thus the waveform of I_(LA) during t_(on)can still be seen as linear and Equation (6) is viable. When thelow-side auxiliary switch LS_(Aux) is on, the expression of I_(LA)during t_(on) is as follows:

$\begin{matrix}{{I_{LA}(t)} = {\frac{V_{out}}{R_{{on}\_ A}}{\left( {1 - {\exp \left( {- \frac{R_{{{on}\_ A}^{t}}}{L_{A}}} \right)}} \right).}}} & (13)\end{matrix}$

and the expressions of capacitor charge from the two output stages are:

$\begin{matrix}{{{Q_{Main} \approx {\frac{1}{2}T_{R}\Delta \; I_{out}}} = \frac{\Delta \; I_{out}^{2}}{2k_{1}}},} & (14) \\{Q_{Aux} = {{\int_{0}^{t_{on}}{{I_{LA}\left( t_{on} \right)}\ {t}}} + \frac{{I_{LA}\left( t_{on} \right)}^{2}}{2k_{3}}}} & (15)\end{matrix}$

By setting

Q _(Main) =Q _(Aux)  (16)

t_(on) can be solved from Equation (16) and t_(off) is calculated as:

$\begin{matrix}{t_{off} = {\frac{I_{LA}\left( t_{on} \right)}{k_{3}}.}} & (17)\end{matrix}$

As will be discussed in greater detail, R_(on) _(—) _(A) must be smallenough so that the resulting t_(on)+t_(off) (auxiliary stage activetime) is shorter than T_(R) (main stage recovery time) under allpossible ΔI_(out). For any given combination of V_(in), V_(out), L_(M)and L_(A), maximum allowable R_(on) _(—) _(A) can be found by letting

t _(on) +t _(off) =T _(R) @ΔI _(out) _(—) _(max).  (18)

Within the range of R_(on) _(—) _(A) larger R_(on) _(—) _(A) results insmaller auxiliary stage transistor but higher output voltage overshoot(in case of heavy-to-light load transient) due to reduced auxiliarycurrent slew-rate. When R_(on) _(—) _(A) is selected by trading-offphysical area and overshoot, actual transistor size can be determinedaccording to the technology used for fabricating the components.

To illustrate the influence of R_(on) _(—) _(A) on the dynamicperformance, a buck converter was designed and simulated in PSIM usingthe parameters in Table I. In steady state operation, the system relieson a traditional linear PWM controller to maintain regulation. Theswitching frequency is 390 kHz. When a load current step occurs, thesystem enters transient recovery mode as the main and auxiliary switchesare controlled using the proposed method. By the end of recovery processthe system returns to linear mode while the output voltage is regulatedby the traditional PWM controller. The main output stage recovery timeand auxiliary output stage turn-on and turn-off time under differentR_(on) _(—) _(A) are calculated with MATLAB and applied to obtain thetransient waveforms.

TABLE I DESIGN PARAMETERS Parameter Value V_(in) 12 V V_(out) 1 V L_(M)2.2 μH L_(A) 820 nH R_(on)_M 25 mΩ R_(on)_A As specified Δ/_(out) −3 to+3 A

FIG. 6 shows the output voltage waveforms during a −3 A load transient.The case with R_(on) _(—) _(A)=0.1Ω has the lowest overshoot (40 mV),but a 20 mV voltage undershoot during transient recovery is observed.For R_(on) _(—) _(A)=0.5Ω, the overshoot jumps up to 50 mV but the totalvoltage deviation is reduced by 10 mV compare to the former case whereno undershoot is observed. In both of these cases, the output voltagesettles immediately after transient recovery, achieving smoothtransition to steady-state linear control. As R_(on) _(—) _(A) increasesto 0.7Ω, the peak overshoot is increased to 58 mV and a secondaryvoltage bump occurs after the linear control takes over, which takesmore than 20 additional switching cycles to settle.

Changes in the output voltage waveform with R_(on) _(—) _(A) can beexplained by analyzing the transient current waveforms in the auxiliaryand main output stages. As indicated in FIG. 7, increasing the value ofR_(on) _(—) _(A) would reduce the slew-rate of auxiliary output stagecurrent I_(LA) and cause it to saturate at a point where the voltagedrop across the low-side auxiliary switch equal to V_(out). Therefore,longer auxiliary output stage active time (the sum of t_(on) andt_(off)) is required to achieve capacitor charge balance.

The voltage overshoot and undershoot during transient recovery mainlydepends on the slew-rate of I_(LA) and the maximum achievable current.When R_(on) _(—) _(A) increases from 0.1Ω to 0.5Ω, the peak auxiliarycurrent drops from over 4 A to 2 A, causing slightly higher overshootwhile eliminating undershoot in the output voltage. The same trendcontinues as R_(on) _(—) _(A) increases to 0.7Ω.

The settling of output voltage after the transition from transientrecovery back to linear control is highly dependent on the main outputstage recovery time and the auxiliary output stage active time. As canbe observed from FIGS. 7-9, for R_(on) _(—) _(A) of 0.1Ω and 0.5Ω, theactive time of the auxiliary output stage is shorter than the recoverytime of the main output stage. Capacitor charge balance is achieved whenthe controller switches back to linear mode. The output voltage reachesits steady-state value at the transition point and settles thereafter.However, for R_(on) _(—) _(A)=0.7Ω, the required active time ofauxiliary output stage is longer than the recovery time for the mainoutput stage and the capacitor charge is not balanced when thetransition happens. After linear control takes over, it tries toregulate the output with the auxiliary output stage still active.Transfer function of the linear control loop is temporarily disturbedsuch that the controller will not function properly until the auxiliaryoutput stage is turned off. This causes a secondary voltage bump and along settling time.

From the above analysis, it can be concluded that the size of the powertransistors in the auxiliary output stage should be selected to ensurethat R_(on) _(—) _(A) does not cause an excessively long auxiliary stageactive time. Based on the knowledge of the maximum possible loadtransient, numerical analysis could be used to estimate the maximumallowable R_(on) _(—) _(A). For the buck converter under study, therequired active times of different auxiliary output stages forheavy-to-light load transient with steps of 0.5 to 3 A are calculatedand annotated in FIG. 10. The calculated recovery time of the mainoutput stage under the same transient condition is also plotted forcomparison. Since the converter is designed to handle load transientwith magnitude up to 3 A, the upper bound of R_(on) _(—) _(A) should beabout 0.5Ω, which means the auxiliary switches can be 20 times smallerthan the main switch given that a 25 mΩ on-resistance main output stageis required to achieve a 93% peak power conversion efficiency. A samplelayout of the dual output stages based on TSMC 0.25 μm 12 V technologyis shown in FIG. 11. It should be noted that the physical size of theoutput stages includes metal connections and isolation guard rings whichdo not shrink proportionally with the actual size of MOSFETs. Thus theauxiliary stage occupies about 8% the total area rather than 4.8% ifonly the transistor size is considered.

While the size of auxiliary output stage must meet the criteriadiscussed above to ensure proper linear controller operation, optimizedsizing for integrated auxiliary switches relies on the trade-offsbetween peak current and power loss in the auxiliary output stage. Smalltransistor size is preferable in terms of lower current peak since lessnumber of bonding wires and I/O pins are needed for chip packaging.However, the increase in on-resistance and lengthened auxiliary stageactive time will lead to a higher power loss during the transientrecovery process. Therefore, heat dissipation should be taken intoaccount for POL converters that undergoes frequent load transient. Forthe buck converter discussed above, increasing R_(on) _(—) _(A) from0.1Ω to 0.5Ω results in 80% smaller size and 50% less current peak, butthe amount of power lost on the auxiliary output stage is increased by2.5 times.

As described above, PID control is suspended during the transientrecovery process, thus the digital controller 16 loses the ability toregulate the duty-ratio with changing load current. When PID control isreactivated by the end of transient recovery, the duty-ratio the digitalcontroller 16 holds is only applicable to the original load currentbefore transient. If the controller starts with the stored duty-ratio,it will cause further output voltage deviation that may even exceed thevoltage peak occurred during transient. This secondary voltage deviationis much more severe for integrated power stages due to the fact thatthey have larger loss elements compare to the discrete implementation.In order to mitigate this problem, the new steady-state duty-ratio afterload transient needs to be predicted and applied to the PID control modeduring the recovery process such that the output voltage settles to nearsteady state immediately after the PID control mode takes over.Accordingly, it is necessary for the digital controller 16 to predictand apply the new steady-state duty-cycle by the end of nonlineartransient recovery to achieve smooth transition back to linear PID mode.

To estimate the difference in duty-cycle before and after each loadtransient, the magnitude of transient step is needed, which is reflectedby the output voltage slew-rate, as expressed below:

$\begin{matrix}{{\Delta \; I_{out}} \approx {C_{out} \cdot {\frac{V_{out}}{t}.}}} & (19)\end{matrix}$

Two consecutive samples of V_(out) are taken by the digital controller16 at the beginning of the transient recovery process with samplinginterval ΔT_(sample), as shown in FIG. 12. Thus Equation (19) isequivalent to:

$\begin{matrix}{{\Delta \; I_{out}} \approx {\frac{1}{C_{out}} \cdot {\frac{\Delta \; V_{out}}{\Delta \; T_{sample}}.}}} & (20)\end{matrix}$

The difference in duty-cycle before and after the load transient is thenestimated as:

$\begin{matrix}{{{\Delta \; D} = {\frac{\Delta \; {I_{out} \cdot R_{loss}}}{V_{in}} \approx {\frac{R_{loss}}{C_{out} \cdot V_{in}} \cdot \frac{\Delta \; V_{out}}{\Delta \; T_{sample}}}}},} & (21)\end{matrix}$

R_(loss) is the equivalent series resistance in the power path.R_(loss)/V_(in) can be calibrated by applying 1 A load current andcalculate as shown below:

$\begin{matrix}{{\frac{R_{loss}}{V_{in}} = {D_{ideal} - D_{I_{out} = {1\; A}}}},} & (22)\end{matrix}$

where D_(ideal) is the nominal V_(out)/V_(in), and D_(lout=1A) is theactual duty-cycle when load current is 1 A. After getting ΔD, thedigital controller 16 applies the new steady-state duty-cycle bysubtracting (in case of heavy-to-light load transient) ΔD from theduty-cycle previously saved.

A fully integrated DC-DC converter with on-chip digitally controlleddual output stages was designed and fabricated in accordance with theinvention. The digital controller incorporated in the DC-DC converterwas designed using TSMC 0.25 μm 2.5V standard cells. The output stagesas well as the associated gate drivers were implemented with TSMC 0.25μm 12V thick oxide devices. Each power transistor was divided into twoidentical segments with the gate driver located in the center. Thislayout strategy reduced the timing mismatch of gate signals to differentfingers of the transistor. The main and auxiliary switching nodes wereimplemented in an interleaving structure so that the power stage currentis evenly distributed. The on-resistance shown in Table II is just anexample of design target. The actual value can vary and/or scaleaccording to different technology and applications. For the prototypeintegrated DC-DC converter implemented, the simulated typical-caseon-resistances of the power transistors under 12V gate-to-source voltage(−12V for PMOS) are shown as below

TABLE II On-Resistances of the Power Transistors Name On-ResistanceHS_(Main)   199 mΩ LS_(Main)    16 mΩ HS_(Aux) 485.1 mΩ LS_(Aux)  90.3mΩ

While the main output stage transistors were designed to achieve around90% peak efficiency, the auxiliary stage transistors were sizedfollowing the criteria discussed in integrated DC-DC converter with anauxiliary output stage for transient suppression”, J. Wang, K. Ng, T.Kawashima, M. Sasaki, H. Nishio, A. Prodić, and W. T. Ng, in Proc.Electron Device and Solid State Circuits, November 2009, pp. 380-383,the content of which is incorporated herein by reference. It can beobserved that the auxiliary output stage imposes less than 20% areaoverhead.

Transient performance of the prototype converter was measured under thetest condition specified in Table III. Transient output voltagewaveforms for a conventional single stage PID controller and theproposed dual-output-stage controller are compared in FIGS. 13 and 14.For a 2.25 A to 0.25 A load transient, a reduction in recovery time from280 μs to 50 μs and in overshoot from 105 mV to 51 mV is observed. Sincethe auxiliary stage is disabled in steady-state, the two control methodsshould have similar efficiency profile.

TABLE III Summary of Test Conditions Parameter Value Vin 6 V (maximum 12V) Vout 1 V LM (main stage) 2.2 μH LA (aux stage) 820 nH Cout 200 μFSwitching 390 kHz Frequency Δ/out Switch from 2.25 A to 0.25 A

The experimental results provided above shows that the present inventioncan reduce the transient overshoot by 50% and recovery time by 80% whilemaintaining similar efficiency when comparing to a conventionalsingle-stage linear converter. Further, the provision of the auxiliaryconverter output stage imposes less than 20% area overhead.

The invention has been described with reference to certain preferredembodiments thereof. It will be understood, however, that modificationsand variations are possible within the scope of the appended claims. Forexample, the digital processor has been described as operating in alinear control mode and a non-linear control mode. It will be understoodthat the functions performed by the digital processor can be implementedutilizing one or more programmable devices or discrete components.

1. A DC-DC converter comprising: a main converter output stage; anauxiliary converter output stage connected in parallel to the mainconverter output stage; and a digital controller coupled to the mainconverter output stage and the auxiliary converter output stage thatoperates in a linear controller mode when no load transient is presentand operates in a non-linear controller mode when a load transient isdetected; wherein the digital controller senses an output voltageslew-rate when the load transient is detected and determines aduty-ratio prediction that is applied in the linear controller mode toensure smooth transition from the nonlinear controller mode back to thelinear controller mode.
 2. The DC-DC converter as claimed in claim 1,wherein the linear controller mode is a proportional-integral-derivativecontroller mode.
 3. The DC-DC converter as claimed in claim 1, whereinthe main converter output stage and the auxiliary converter output stageeach include a high side switch, a low side switch and an inductor. 4.The DC-DC converter as claimed in claim 1, wherein the digitalcontroller disables the auxiliary converter output stage when no loadtransient is detected and the DC-DC converter is operated in a steadystate condition.
 5. The DC-DC converter as claimed in claim 1, whereinan on-resistance of auxiliary converter output stage power transistorsprovided in the auxiliary converter output stage is such that theauxiliary converter output stage active time is less than the mainconverter stage recovery time.
 6. The DC-DC converter as claimed inclaim 5, wherein the upper bound of the on-resistance is the upper boundis about 0.5Ω.
 7. The DC-DC converter as claimed in claim 2, wherein theratio of inductance of the main converter output stage to the auxiliaryconverter outputs stage (L_(M)/L_(A)) is such that, during transientrecovery, the induced output voltage undershoot ΔV_(Undershoot) isalways less than or equal to the output voltage ΔV_(Overshoot) overshoot8. A method of operating a DC-DC converter that includes an auxiliaryconverter output stage connected in parallel to the main converteroutput stage, and a digital controller coupled to the main converteroutput stage and the auxiliary converter output stage that operates in alinear controller mode when no load transient is present and operates ina non-linear controller mode when a load transient is detected, themethod including: operating the digital controller in a linear controlmode when no load transient is detected by the digital controller inorder to maintain the DC-DC converter in a steady state condition;activating the digital controller to operate in a non-linear controlmode when a load transient is detected by the digital controller;performing load-step sensing using the digital controller; generatingswitching commands for the main converter output stage and the auxiliaryconverter output stage with the digital controller based on theload-step sensing; performing a duty-ratio prediction with the digitalcontroller and applying it to the linear control mode to ensure a smoothtransition from the non-linear control mode to the linear control mode.9. The method of operating a DC-DC converter as claimed in claim 7,further comprising introducing a blocking state until the output voltageof the converter settles to steady state.
 10. The method of operating aDC-DC converter as claimed in claim 7, wherein the linear control modeis a proportional-integral-derivative control mode.
 11. The method ofoperating a DC-DC converter as claimed in claim 7, wherein the ratio ofinductance of the main converter output stage to the auxiliary converteroutputs stage (L_(M)/L_(A)) is such that, during transient recovery, theinduced output voltage undershoot ΔV_(Undershoot) is always less than orequal to the output voltage ΔV_(Overshoot) overshoot